Dual-Source Single-Inductor 0.18-μm CMOS Charger–Supply with Nested Hysteretic and Adaptive On-Time PWM Control

Since fuel cells store more energy and batteries supply more power, fuel cell–battery systems can be smaller than single-source supplies. The 0.18-μm CMOS switched-inductor charger–supply shown, for example, draws constant power from an energy source and supplementary power from a battery to supply a 0.8-V load and recharge the battery with excess power. With 62% – 83% efficiency across 0.1 – 8-mA and ±40 mV of worst-case ripple, the system requires 65% less space than a single source occupies. A major challenge with emerging microsensors, biomedical implants, and other portable devices is operational life, because tiny batteries exhaust quickly. And even though 1-g fuel cells store 5× – 10× more energy than 1-g Li Ions, fuel cells supply 10× – 20× less power [1]. This means fuel cells last longer with light loads and Li Ions output more power across shorter periods. So when peak power is far greater than average power, which is typically the case in wireless sensors, for example, a hybrid can occupy less space than one source [1]. Still, managing a fuel cell and a battery to supply a load and recharge the battery, which also acts as an output, with little space is difficult. Switched-inductor circuits are appealing in this respect because they draw and supply more power with higher efficiency than their linear and switched-capacitor counterparts. Inductors, however, are bulky, so microsystems can only rely on one inductor [2–3]. Today, most single-inductor multipleoutput systems derive power from one source [3–4, 6], so the fuel cell and battery require considerable space. And those that use two sources [5] do not manage how much power each should supply across loading conditions, and while [2] does, the efficiency of [2] is low. The advantage of the prototyped 0.18-μm CMOS dual-source single-inductor system built and presented here is less overall volume because it incorporates the functional intelligence of [2] with much higher efficiency. I. Operational Modes Because the 1.1 – 1.3-V energy source vES supplies more energy when delivering constant power, the prototyped system in Figs. 1 and 7 draws constant power PES from vES and supplementary power from the 1.8-V power source vPS to supply a 8mA, 0.8-V load. When PES exceeds the needs of the load in PO, the system uses the excess to recharge vPS. Since the hybrid supply uses and switches a 50-μH 6 × 6 × 2-mm3 inductor LO to transfer power between vES, vPS, and the output vO, the purpose of capacitors CIN and CO is to suppress switching noise in vES and vO. This way, when PES surpasses PO, CO's voltage and vO rise above the reference VREF to such an extent that comparator CPM trips to push the system into the lightload region. CPM pulls the system back into the heavy-mode region when the opposite happens, when PO exceeds PES to pull vO below CPM's lower hysteretic threshold. II. Light-load Region When lightly loaded, comparator CPLT regulates vO about VREF and LO conducts in discontinuous conduction mode (DCM) across the period of the 40-kHz clock fCLK. More specifically, CPLT senses vO to determine which output: vO or vPS, should receive LO's energy. For this, SES and SE in Figs. 1 and 2 first energize LO from vES to ground across τEN's 1.2-μs pulse width to raise LO's current iL from zero to 30 mA. Afterwards, SES and SE open and vE.OFF in Fig. 2 rises to close SDE and either SO or SPCHG. If comparator CPLT senses vO is below VREF by 10 mV, SO drains LO into vO; otherwise, SPCHG depletes LO into vPS. Comparators CPIOZ and CPIPZ then disengage SO and SPCHG together with SDE when SO's and SPCHG's current iL nears zero, when LO is close to empty, which happens at 2.7 and 27 μs in Fig. 2. All switches remain open after that until fCLK initiates another cycle. Luckily, CPLT, CPIOZ, and CPIPZ need not operate across fCLK's entire 25-μs period. CPLT, for one, needs to sense vO only at the end of τEN's 1.2-μs pulse width. This is why fCLK in Fig. 2 engages CPLT a short delay τD after τEN rises, to be ready by the end of τEN, and disengages CPLT another short delay τD after τEN falls. Similarly, CPIOZ and CPIPZ must sense only when SO and SPCHG conduct iL, so CPLT's output vLT enables CPIOZ and CPIPZ and CPIOZ's and CPIPZ's flip flops disable them after they detect iL nears zero. Duty-cycling CPLT, CPIOZ, and CPIPZ this way reduces their power consumption by 90%. II. Heavy-load Region When heavily loaded, LO draws one energy packet from vES and one variable packet from vPS that transconductor GHV in Figs. 1 and 3 controls when regulating vO about VREF. As with light loads, LO stops conducting after that until fCLK starts another cycle. Comparator CPHV compares GHV's slow-moving output vG against a triangular saw-tooth voltage vSAW to pulse-width modulate (PWM) how long LO energizes from vPS. For all this, like before, SES and SE first energize LO from vES to ground across τEN's 1.2-μs pulse width to raise LO's current iL from zero to 30 mA. Afterwards, SES and SE open and SDE and SO close to drain LO into vO. SDE then opens and, if GHV senses that vO still needs power, vSAW starts ramping and SPE closes to energize LO from vPS to vO. When vSAW falls below GHV's vG, SPE opens and SDE closes to deplete LO into vO. SDE and SO then open when CPIOZ in Fig. 3 senses that LO's iL is nearly zero, after which point all other switches remain open until the next fCLK cycle. III. Measured Performance As Fig. 4 illustrates, vO ripples at ±2.5 mV when lightly loaded with 0.1 mA and ±40 mV when heavily loaded with 8 mA. The ripple is higher at 8 mA because vES and vPS deliver power early in the period and the load slews CO afterwards, when disconnected from LO. Since CPM determines which mode to assert in hysteretic fashion, the system transitions through modes across rising and falling 0.1 – 8-mA load dumps quickly and without ringing oscillations. When the load is light at 0.1 – 1 mA, the fraction of vES power that vO and vPS receive is 62% – 73%, as Fig. 5 shows. And the fraction of power vO receives from vES and vPS when heavily loaded with 1 – 8 mA is 62% – 84%. Power-conversion efficiency ηC bottoms when the system transitions across 1 mA and peaks to 73% under hysteretic control below 1 mA and 84% under PWM control above 1 mA. ηC peaks at two points because switches are smaller in light mode than in heavy mode, so conduction and gate-drive losses balance at two load levels. IV. Conclusions The key feature of the single-inductor 0.18-μm CMOS charge–supply prototyped and validated here is managing two complementary sources with 62% – 84% power-conversion efficiency. For this, the system duty-cycles circuit blocks, operates the inductor in discontinuous conduction mode, and employs hysteretic and PWM control schemes to regulate the output in and across light and heavy modes. The challenge with single-source systems when lightly loaded over extended periods and pulsed periodically with heavy loads, as in the case of wireless sensors, is that oversizing a fuel cell to output more power or a Li Ion to last longer demands more space than an efficient hybrid. To sustain a 0.1 – 10-mW load for one month, for example, [4] and [6] in the table of Fig. 6 require a 1-g fuel cell to supply 10 mW or a 0.45-g or 0.43-g Li Ion to last one month. And because [5] cannot adjust how much power each source should supply according to the load, [5] similarly needs a 1-g fuel cell or a 0.43-g Li Ion. [2] can manage a fuel cell and a Li Ion according to the load, but the cost of intelligence, robustness, and accuracy is unfortunately efficiency, so this hybrid system demands more space than [4–6]. The dual-source single-inductor charger–supply presented here, however, requires a 0.1-g fuel cell and a 0.05-g Li Ion,which combined is 65% less weight at 0.15 g than that of the smallest counterpart.AcknowledgementThe authors thank Pooya Forghani, Paul Emerson, and Texas Instruments for their support and for fabricating theprototyped IC. References[1] M. Chen, J.P. Vogt, and G.A. Rincón-Mora, "Design Methodology of a Hybrid Micro-Scale Fuel Cell-Thin-Film LithiumIon Source,” IEEE International Midwest Symp. Circuits and Systems, pp. 674-677, Aug. 2007.[2] S. Kim and G.A. Rincón-Mora, "Single-Inductor Dual-Input Dual-Output Buck-Boost Fuel Cell-Li Ion Charging DC-DCConverter," IEEE International Solid-State Circuits Conference (ISSCC), pp. 444-445, San Francisco, CA, Feb. 2009. [3] D. Ma, W.H. Ki, C.Y. Tsui, and P.K.T. Mok, “Single-Inductor Multiple-Output Switching Converters with Time-MultiplexingControl in Discontinuous Conduction Mode,” IEEE J. Solid-State Circuits, vol. 38, no. 1, pp. 89-100, Jan. 2003.[4] M.H. Huang and K.H. Chen, “Single-Inductor Multi-Output (SIMO) DC-DC Converters with High Light-Load Efficiencyand Minimized Cross-Regulation for Portable Devices,” IEEE J. Solid-State Circuits, vol. 44, no. 4, pp. 1099-1111, Apr. 2009.[5] K.W.R. Chew, Z. Sun, H. Tang, and L. Siek, “A 400nW Single-Inductor Dual-Input-Tri-Output DC-DC Buck-BoostConverter with Maximum Power Point Tracking for Indoor Photovoltaic Energy Harvesting,” ISSCC Dig. Tech. Papers, pp.68-69, Feb. 2013.[6] Y. Qiu, C.V. Liempd, B. Op het Veld, P.G. Blanken, and C.V. Hoof, “5μW-to-10mW Input Power Range Inductive BoostConverter for Indoor Photovoltaic Energy Harvesting with Integrated Maximum Power Point Tracking Algorithm,” ISSCC Dig.Tech. Papers, pp. 118-120, Feb. 2011. Captions:Figure 1: Dual-source single-inductor 0.18-μm CMOS charger–supply. Figure 2: Light-load circuit and related waveforms. Figure 3. Heavy-load circuit and related waveforms.Figure 4: Rising and falling load-dump responses and output regulation across operating modes.Figure 5: Simulated and measured power-conversion efficiency. Figure 6: Performance summary and comparison with the state of the art.Figure 7: Fabricated die and experimental printed-circuit board. Figure 1: Dual-source single-inductor 0.18-μm CMOS charger–supply. Figure 2: Light-load circuit and related waveforms. Figure 3: Heavy-load circuit and related waveforms.